Errata
Below are a listing of known errors in the book’s first printing.
Page No.  Section  Example/ Problem  Figure  Equation reference  Original  Correction 
23  1.2.3  1.8  none  none  Using (1.67), we find for V_{ds} = V_{eff} = 0.4V, 
Using (1.67), we find for V_{ds} = V_{eff} = 0.35V, 
23  1.2.3  1.8  none  none  In the case where V_{DS} = V_{eff} + 0.5V = 0.9V, we have  In the case where V_{DS} = V_{eff} + 0.5V = 0.85V, we have 
33  1.2.7  1.13  none  none  C_{ov} = 2.0 x 10^{4} pF/μm  C_{ov}‘ = 2.0 x 10^{4} pF/μm 
47  1.4.2  none  1.31  none  In the plot of I_{d} vs. V_{gs}, the slope of I_{d} vs. V_{gs} in mobility degradation looks same as or larger than that of square law, which contradicts the plot of g_{m} vs. V_{gs} 
Eliminate curvature on I_{d} vs V_{gs} plot in mobility degradation. 
53  1.5.3  none  1.33  none  Two text labels include the “section end” character 
Replace with “/” 
62  1.7.1  none  none  1.146  D_{n}  D_{p} 
70  1.10.2  P1.12  none  none  The technology node and the values of V_{db}, V_{sb}, and C_{jsw} are not given 
Assume the 0.35μm CMOS technology, V_{db} = V_{sb} = 0, and ignore sidewall capacitances. 
101  2.3.3  none  none  none  g_{m}/I_{D} = 1/(2V_{eff})  g_{m}/I_{D} = 2/V_{eff} 
102  2.3.3  none  none  2.19  σ^{2}(ΔV_{GS}) = 1/WL[A_{Vt0}^{2}(g_{m}/I_{D})^{2} + A_{K’}^{2}]  σ^{2}(ΔV_{GS}) = 1/WL[A_{Vt0}^{2} + (I_{D}/g_{m})^{2}A_{K’}^{2}] 
106  2.4.2  2.6  2.31  none  left and right sides of the figure have been cropped 
left and right layout labels should indicate 10μm and 19.6μm dimensions respectively 
124  3.4  none  none  none  …, the input impedance, r_{out}, is found to be 1/g_{m}…  …, the input impedance, r_{in}, is found to be 1/g_{m}… 
140  3.11.2  P3.12  none  none  … shown in Fig. P3.11. 
… shown in Fig. P3.12. 
145  4.1  none  none  none  Further, the real parts of all the poles will be positive…  Further, the real parts of all the poles will be negative… 
148  4.1.2  4.2  none  none  … for the case f = 10 Hz, where ω = 2πf = 628.3 rad, …  … for the case f = 10 Hz, where ω = 2πf = 62.83 rad, … 
154  4.1.2  4.5  none  4.60  ∠A(ω) = tan(ω/ω_{3dB})^{1}  ∠A(ω) = tan^{1}(ω/ω_{3dB}) 
155  4.1.3  none  none  4.70  Φ = tan^{1}(ω/ω_{p1}) – tan(ω/ω_{p2})  Φ = tan^{1}(ω/ω_{p1}) – tan^{1} (ω/ω_{p2}) 
161  4.1.4  4.7  none  4.93  1.05 x 10^{8} rad/sec  1.05 x 10^{10} rad/sec 
161  4.1.4  4.7  none  4.94  5 x 10^{6} rad/sec  5 x 10^{8} rad/sec 
170  4.2.3  4.8  none  none  … to the case where Y(s) = 1/sC.  … to the case where Y(s) = sC. 
175  4.2.4  4.9  4.19  none  g_{m1} labels should be beside the dependent current sources 

176  4.2.4  4.10  none  none  use the transistor parameters listed in Table 1.5 for the 0.8μm CMOS technology in Table 1.5. 
use the transistor parameters listed in Table 1.5 for the 0.8μm CMOS technology. 
176  4.2.4  4.10  none  4.158  the denominator of equation (4.152):R_{in}[C_{gs1} + C_{gs1}(1 + g_{m1}R_{2})] + R_{2}(C_{gd1} + C_{2}) 
the denominator of equation (4.152):R_{s}[C_{gs1} + C_{gs1}(1 + g_{m1}R_{2})] + R_{2}(C_{gd1} + C_{2}) 
177  4.2.5  4.11  4.13  none  L_{1}=A_{o} l L_{1} V_{eff,1}  L_{1}=(A_{o} l L_{1} V_{eff,1}) / 2 
179  4.2.5  4.12  none  none  2 x 355mA x 0.18μm / (270mA/V^{2} x (0.1V)^{2}) 
2 x 350mA x 0.18μm / (270mA/V^{2} x (0.1V)^{2}) 
191  4.4  4.16  none  4.212  = (4.27 x 10^{9} rad/s = 2π x 680 MHz) 
= 4.27 x 10^{9} rad/s = 2π x 680 MHz 
194  4.5.1  none  4.31  none  capacitor symbol overlaps the label, C_{db} 
move capacitor symbol to the left 
195  4.5.3  none  none  4.226  negative sign in the numerator  no negative sign in the numerator 
196  4.5.3  none  4.34(a)  none  label “V_{out}” is mislocated  move “V_{out}” label to the output below “R_{D}“ 
196  5.4.3  none  4.35  none  dependent current source on the right points down  dependent current source on the right points up 
197  5.4.3  none  4.36  none  dependent current source on the right points down  dependent current source on the right points up 
197  5.4.3  none  4.37  none  dependent current source on the right points down  dependent current source on the right points up 
205  5.1.2  none  none  5.7  ΔA / [(1+L)^{2}+ ΔAβ]  ΔA / [(1+L)^{2} + ΔAβ(1+L)] 
209  5.2  5.3  none  none  In practice, the voltage at v_{–} will increase above v_{in}  In practice, the voltage at v_{+} will increase above v_{in} 
210  5.2.1  none  none  none  Key Point: “… 180 degres …” 
Key Point: “… 180 degrees …” 
215  5.3.1  none  none  5.29  Second equals sign: “=” 
Second equals sign should be approximately equals 
216  5.3.1  5.6  5.9(b)  none  expression in the figure: “s/ω_{ta}“ 
expression in the figure: ” ω_{ta}/s” 
217  5.3.1  5.6  none  none  … is well approximated by s/ω_{ta}.  … is well approximated by ω_{ta}/s. 
217  5.3.1  5.6  none  5.32  first term on right side: “(s/ ω_{ta})” 
first term on right side: “(ω_{ta} /s)” 
221  5.4  none  none  none  Key Point: “… desired gain, β …” 
Key Point: “… desired gain, 1/ β …” 
225  5.4.1  5.9  5.13(b)  none  arrow points down on the dependent source g_{m,1}  arrow points up on the dependent source g_{m,1} 
225  5.4.1  5.9  5.13(c)  none  line connecting the tops of Z_{x} and C_{gs,5} is broken  line connecting the tops of Z_{x} and C_{gs,5} should be connected 
225  5.4.1  5.9  5.13(c)  none  C_{gs,1/2}  C_{gs,1} /2 
229  5.4.2  5.11  none  none  2π 2.86MHz  2π 28.6MHz 
235  5.5  none  none  none  Key Point: “… 180 degres …” 
Key Point: “… 180 degrees …” 
235  5.5  none  none  none  Key Point: “… desired gain, β…” 
Key Point: “… desired gain, 1/β …” 
244  6.1.1  6.1  none  none  We can calculate the transconductances of Q_{1}, Q_{2}, Q_{7} and Q_{8}. 
We can calculate the transconductances of Q_{1}, Q_{2}, and Q_{7}. 
245  6.1.1  6.1  6.3  none  A_{v1} A_{v2} = 1043 V/V or 60.4dB 
A_{v1} A_{v2} = 475 V/V or 53.5dB 
248  6.1.2  6.3  none  unnumbered  ω_{p2} = 3.12mA/V / (5pF + 1pF) = 2π 82.8 MHz  ω_{p2} = 3.12mA/V / 5pF = 2π 99.3 MHz 
249  6.1.2  6.3  none  unnumbered  ω_{p2} = 3.12mA/V / 5pF = 2π 99.3 MHz  ω_{p2} = 3.12mA/V / 1pF = 2π 497 MHz 
254  6.2.1  none  none  none  …introduce an additional approximately +20° phase shift…  …introduce an additional approximately +20° to 30° phase shift… 
258  6.2.2  6.7  none  none  the opamp and ω_{z}, ω_{p}, and ω_{2} are the frequencies of a zero, the dominant pole, and the equivalent second pole, respectively. 
the opamp and ω_{z}, ω_{p1}, and ω_{2} are the frequencies of a zero, the dominant pole, and the equivalent second pole, respectively. 
259  6.2.3  none  none  none  Repeating equations (6.9) and (6.20) here, we have  Repeating equations (6.42) with ß=1 and(6.20), we have 
261  6.2.3  none  none  6.71  lefthand side, denominator: V_{eff16}  lefthand side, denominator: V_{eff9} 
268  6.4  none  6.19  none  Dependent current source, G_{ma}v_{i} points down  Dependent current source, G_{ma}v_{i} points up 
279  6.5  6.11  none  none  the unitygain frequency is increased by 150% 
the unitygain frequency is increased by 100% 
292  6.8  none  6.38  none  Grounded nodes in the schematic  These are AC grounds, and should be connected to a commonmode reference voltage 
304  7.1.1  7.1(b)  none  none  V_{b} = 1.1(450mV)+(200mV)/(0.9 √0.9) = 729mV  V_{b} = 1.1(450mV)+(200mV)/(√0.9 √0.9) = 717mV 
307  7.2.1  none  7.6  none  Q_{13} size is 100 
Q_{13} size is 25 
309  7.2.1  none  none  none  effective gatesource voltages increase by 27 percent  effective gatesource voltages increase by 39 percent 
309  7.2.2  none  none  none  Q_{13} is five times greater … gate voltage of Q_{13} …  Q_{14} is five times greater … gate voltage of Q_{14} … 
310  7.2.2  none  7.8  none  Resistor label “RB” 
Resistor label should use subscript: “R_{B}“ 
311  7.3.1  none  none  none  Vbe will have approximately a 2mV/°K  Vbe will have approximately a 2mV/K 
313  7.3.1  7.4  none  none  For the special case of T_{0} = 300°K 
For the special case of T_{0} = 300K 
313  7.3.1  7.4  none  none  at 300°K  at 300K 
313  7.3.1  7.4  none  none  Recalling that 0°K  Recalling that 0K 
314  7.3.1  7.4  none  7.29  8 mV/°K / 1.24V = 6.5×10^{6} parts/°K = 6.5ppm/K  8 mV/K / 1.24V = 6.5×10^{6} parts/K = 6.5ppm/K 
314  7.3.2  none  none  none  the same collector currents and collectoremitter voltages  the same collector currents and collectorbase voltages 
317  7.3.2  none  7.13(b)  none  opamp positive terminal on top  opamp positive terminal on bottom 
318  7.3.2  7.5  none  7.51  Now, recalling from (7.26) that 
Now, recalling from (7.25) that 
320  7.3.3  none  7.15  none  Leftmost resistor is R_{a}  Leftmost resistor is R_{a}/M 
320  7.3.3  none  7.15  none  Polarity of the opamp is incorrect with the positive terminal on the left  Switch the polarity of the opamp so that positive terminal is on the right 
329  7.7.2  P7.7  none  none  Assume (W/L)12,13 = (2μm/0.2μm) and (W/L)12,13 = (8μm/0.2μm).  Assume (W/L)_{14} = (2μm/0.2μm) and (W/L)_{15} = (8μm/0.2μm). 
329  7.7.2  P7.10  none  none  an dc input  a dc input 
329  7.7.3  P7.12 and P7.13  none  none  320°K  320K 
330  7.7.4  P7.20(d)  none  none  … so that ω_{pL} ≈ ω_{pa}/100? 
… so that ω_{pL} ≈ ω_{pa}/10? 
361  8.9.1  8.2  P8.2  none  V_{dd}=5V  V_{cc}=5V 
361  8.9.1  8.9  P8.9  none  V_{dd}=5V  V_{cc}=5V 
371  9.2.3  9.4  none  none  K_{v}^{2}=(50nV^{2})  K_{v}^{2}=(50nV)^{2} 
378  9.3.1  none  none  none  temperature (300°K)  temperature (300K) 
383  9.3.6  9.8  none  none  temperature of 300°K  temperature of 300K 
392  9.4.2  9.11  none  none  i.e. 300°K  i.e. 300K 
400  9.5.2  none  none  none  Δω = ω_{2} – ω  Δω = ω_{2} – ω_{1} 
403  9.5.3  9.14  none  none  ID3 = 45 dB  ID3 = 45 dB 
408  9.8.2  P9.11 and P9.12  P9.11 and P9.12  none  yaxis title: V_{rms}  yaxis title: V_{rms}^{2} 
420  10.3  10.3  none  none  V_{DD} = 1V  V_{DD} = 2V 
420  10.3  10.3  none  none  2.8mV charge injection; total of 56mV  28mV charge injection; total of 80mV 
467  11.9.3  P11.11  none  none  Equation denominator: “C_{hld} + C_{be6}“ 
Equation denominator: “C_{hld} + C_{be6} + C_{F}“ 
471  12.1.2  none  none  none  independent of the feedin gain, K_{1,2,3}.  independent of the feedin gain, K_{0,1,2}. 
471  12.1.2  none  none  none  determined by the feedin gain, K_{1,2,3}.  determined by the feedin gain, K_{0,1,2}. 
472  12.1.2  12.1  12.3  none  1/Q = √2  1/Q = √2 
477  12.2.4  none  none  12.26  s(G_{m3} / C_{x}+ C_{b})  s(G_{m3} / C_{x} +C_{B}) 
478  12.2.4  none  none  12.31  G_{m3} / C_{x}+ C_{b}  G_{m3} / C_{x} +C_{B} 
482  12.3  12.5  none  none  For the case r_{e,max} < 0.01R_{e}  For the case r_{e,max} < 0.01R_{s} 
482  12.3  none  12.13  none  G_{m} = 1/R_{e}  G_{m}= 1/R_{s} 
482  12.3  none  12.14  none  G_{m} = 1/R_{e}  G_{m} = 1/R_{s} 
510  12.8.1  12.10  12.41(b)  none  V_{no}  V_{po} 
510  12.8.1  12.10  12.41(b)  none  V_{po}  V_{no} 
536  12.13.4  P12.20  none  none  Sketch the resulting magnitude response V_{in}(ω)/V_{out}(ω) … 
Sketch the resulting magnitude response V_{out}(ω)/V_{in}(ω) … 
554  13.6  13.8  none  none  at 40kHz and 60kHz is 0.2399 and 0.1559, 
at 40kHz and 60kHz is 0.2339T and 0.1559T, 
568  14.2.3  none  14.13(a)  none  v_{co}(nTT)  v_{co}(nTT/2) 
568  14.2.3  none  14.13(b)  none  v_{ci}(nTT/2) and v_{co}(nTT/2)  v_{ci}(nT) and v_{co}(nT) 
571  14.3  none  none  none  … a rms voltage V_{c1}^{2}_{(rms)}= ÖkT/C …  … a rms voltage V_{c1 (rms)}= ÖkT/C … 
571  14.3  none  none  none  … capacitor noise voltage V_{c2}^{2}_{(rms)}= ÖkT/C. 
… capacitor noise voltage V_{c2(rms)}= ÖkT/C. 
573  14.4  14.3  none  none  the bilinear transform p = (z1) / (z+1) 
the bilinear transform s = (z1) / (z+1) 
574  14.4  none  none  none  As in Example 14.2,  As in Example 14.3, 
577  14.4.2  none  14.21  none  Φ_{1}, Φ_{2}  swap clock phases near C_{3}: Φ_{2}, Φ_{1 } 
605  14.12.6  P14.20  none  none  ensure a dc offset of less than 5mV 
ensure a output dc offset of less than 5mV 
605  14.12.6  P14.21  none  none  Find the resulting dc offset 
Find the resulting output dc offset 
619  15.5.3  15.4  none  none  For the gain error, from (15.25) we have 
For the gain error, from (15.24) we have 
627  16.1.2  none  16.4  none  The LSB decoder at the bottom of the figure must also accept as input the digital signal b_{2} since the folded resistorstring changes direction.  
642  16.5  none  none  none  … employ an array of 2N1 unitary … 
… employ an array of 2^{N}1 unitary … 
669  17.4.1  17.7  none  none  V_{3} = 2(V_{2}V_{ref}/4) = 540V 
V_{3} = 2(V_{2}V_{ref}/4) = 540mV 
669  17.4.1  17.7  none  none  V_{in} = 375mV 
V_{in} = 312.5mV 
669  17.4.2  none  none  17.42  V_{in} = V_{ref}/2 (S(b_{i,0 }+ b_{i,1 }– 1)2^{i} + b_{n1,1}2^{N2} +b_{N1,0}2^{N1}) 
V_{in} = V_{ref}/2 (S(b_{i,0} +b_{i,1 }– 1)2^{i} + b_{n1,1}2^{N+2} +b_{N1,0}2^{N+1}) 
671  17.4.2  17.8  none  none  V_{2} = 2(V_{1} – V_{1x}) = 200mV 
V_{2} = 2(V_{1} – V_{1,x}) = 200mV 
671  17.4.2  17.8  none  none  leftalign the variables “b_{1,0}“, “b_{1,1}“, and “b_{2,1} b_{2,0}” in the table 

681  17.7  none  17.31  none  The bottom of the resistor string is grounded.  The bottom of the resistor string should be connected to an additional amplifier output that compares the input, V_{in}, to ground. 
693  17.12.5  P17.22  none  none  the reference voltage supply V_{ref}.  the reference voltage supply V_{ref} = 0.5V. 
711  18.3.1  none  none  none  f_{0} = 44.1 kHz  2f_{0} = 44.1 kHz 
720  18.5.2  none  none  18.44  RHS of 1st line: Ŝ_{TF2}(z)X_{1}(z) + N̂_{TF1}(z)X_{2}(z)  Ŝ_{TF2}(z)X_{1}(z) – N̂_{TF1}(z)X_{2}(z) 
720  18.5.2  none  none  18.44  RHS of 2nd line: all terms are positive  RHS of 2nd line: 3rd and 4th terms (those including N̂_{TF1}) should be negative 
720  18.5.2  none  none  18.45  RHS: Ŝ_{TF2}(z)S_{TF1}(z)E_{1}(z) + N̂_{TF1}(z)N_{TF2}(z)E_{2}(z)  Ŝ_{TF2}(z)S_{TF1}(z)E_{1}(z) – N̂_{TF1}(z)N_{TF2}(z)E_{2}(z) 
722  18.6  none  18.23  none  Modulator as drawn is unstable 
Modify H(z) so that z^{2} appears in the feedback loop and there is no delay in the feedforward path. 
729  18.8.4  none  none  18.55  first term on the righthandside of the equals sign: U(z)z^{1}  U(z)z^{1}/(2z^{1}) 
729  18.8.4  none  none  18.55  exponent in the 3rd and 4th terms on the righthand side: (1z^{1})^{2} 
(1z^{1})^{2} 
730  18.8.4  none  18.29  none  Gain “1/2” appears between 2nd and 3rd summers at the top left of the figure 
To make it consistent with the reference [Hairapetian, 1994], the gain “1/2” should be moved to between 1st and 2nd summers at the top left of the figure 
730  18.8.4  none  18.29  none  box at lowerright hand corner, with U_{s}(z) as its input, contradicts the reference [Hairapetian, 1994]: 2(1z^{1})^{2} 
2z^{1} 
742  19.1.3  19.1  19.3 and 19.4  none  waveform labels: V_{osc}  waveform labels: V_{div} 
750  19.2.1  none  19.9  none  caption: “(a) timedomain step response; (b) frequency domain jitter transfer function magnitude response”  caption: “(a) frequency domain jitter transfer function magnitude response; (b) timedomain step response” 
760  19.3  19.11  none  none  1st and 3rd equations, second bracketed term includes “g_{m}“ 
1st and 3rd equations, second bracketed term, “g_{m1}“ 
760  19.3  19.11  none  none  2nd equation, righthand side: “n_{k} / V_{s}” 
2nd equation, RHS: “n_{k} / V_{s} g_{m} (R_{d}r_{o1})” 
760  19.3  19.11  none  none  3rd equation, term preceding the brackets: “(R_{d}r_{o1})^{2} / V_{s}^{2}“ 
3rd equation, term preceding the brackets: “1 / g_{m}^{2} V_{s}^{2}“ 
778  19.5.2  19.19  none  19.95  denominator is incorrect 

778  19.5.2  19.19  none  19.96  denominator: “2ω_{pll}“  denominator: “4ω_{pll} “ 
784  19.8.3  P19.14  none  none  zero crossing times: “… 40.2ns, 41.0ns, 49.0 ns, …” 
zero crossing times: “… 40.2ns, 49.0 ns, …” 
End papers  none  none  none  none  g = 2I_{D} / V_{eff}  g_{m} = 2I_{D} / V_{eff} 
End papers  none  none  none  none  Endpaper CMOS technology parameter values differ slightly from those in Table 1.5. 
Posted problem solutions use the values in Table 1.5. 
Section 18.3.1 (p.710) specifies fs=5.6448 MHz and f0=44.1 kHz to give an OSR=128. But per the definition of OSR used in the book (eq. 18.8), this should give an OSR=fs/(2*f0)=64. I assume the intention here might be to say 2*f0=44.1 kHz ?
Yes you are correct – thanks for the comment. I will add it to the errata.
Section 7.3.3, Fig. 7.15, and Eq. (7.57):
The parameter “N” is not defined in the text.
Yes, this parameter is not explicitly defined. It is hopefully obvious to the reader that “N” refers to the number of diodes in parallel.
There is an error in the errata. For the correction of the mistake in page 320 regarding opamp polarity, it is mentioned that the ref figure is “none”. The ref figure is Fig. 7.15. Please correct.
Fixed – thanks!
Page 314, Section 7.3.2:
“both transistors have the same collector currents and collectoremitter voltages”
Should be “and collectorbase voltages”.
Agreed – I have added this to the errata.
Section 7.2.1, page 309, 2nd paragraph:
1) The reduction in mobility is 28%, not 27%.
2) If the mobility decreases by 28%, then Veff should be: Veff/0.72 = 1.39Veff –> Veff should increase by 39%, not 27%.
1) My calculation gave 0.2786 – a minor point!
2) Agreed – Veff should increase by 39%, not 27%. I have added this to the errata – thanks!
Example 7.1b)
ID is in inverse proportion to R.
BUT Veff is proportional to sqrt of ID rather than ID.
Thus, “0.9” should be replaced by “sqrt(0.9)”.
The correct answer should be 717mV
Yes – thank you! I’ve put this on the errata.
In Section 7.2.2, second paragraph, third line, TWO occurrences:
Q13 should be replaced by Q14.
Right. The current density of Q14 is five times greater than that of Q4. Hence, the gate voltage of Q14 is suitable for use as the PMOS cascode bias Vcascp. Added to the errata – thank you!
page 420,Example 10.3, solution part:second line,”using (10.8),this …=2.8mV”,i think it is “28mV”,dose it mean: 2Vx1.4fF/101.4fF=2.8mV ?
Yes – you are correct. The charge injection due to overlap capacitance should be 27.6mV and so the total charge is found by adding 27.6mV and 52.7mV for a total of 80mV. I have added this to the errata.
On page 24(second ed.), you state that for a pchannel MOS to conduct, it must have VSG>Vtp, where Vtp is a negative quantity. So as a consequence, VSG=0 gives a conducting transistor since 0 is greater than any negative number, but this cannot be true ??
It is easy to get the signs confused in this context. Without having to get into the weeds of it myself, it is clear that your understanding of the matter is correct!
Dear Professors,
The equations 5.2 and 5.3 in Chapter 5 should have a negative sign for the loopgain term. If we set input to zero for loopgain calculations, and follow the loop in Fig.5.1, v=ABx and L=AB, due to terminal v multiplying with 1. This also implies that there is a negative feedback. These equations should be included in the list of errata.
This book is one of my favorites and I liked this time in 2nd edition, topic of feedback was given more attention.
Thanks
The book adopts the more common convention of defining the loop gain of a negative feedback amplifier as positive (at low frequency). The signal x experiences gains of A and then B to the point V so the gain from X to V is AB not AB. If one defines L=AB, then all the equations of 5.4 and onward must also be modified.
V=ABX=AB(UV) so V=U.AB/(1+AB) & Y=A(UV) so Y=U.A/(1+AB)
In the solution of Example 5.1 on page 206 the change in the closedloop gain should be less than 5%, because 4.97514.9975=0.0224, which is 2.24% change.
The change in gain is expressed as a percentage of the originally computed gain. Hence,
4.97514.9975/4.9975 * 100 = 0.00448*100 = 0.45%
I believe 4.133 is incorrect. It should be
wp2 = 1/(RS*(Cgs1+Cgd1))
4.133 is derived assuming C2 is large directly on 4.127,
However we cannot not use 4.127 directly (4.127 is based on the assumption that Cdg1*gm1*R2 >> R2*C2 which is in contradiction with the case we are considering in 4.132.
There is no contradiction in the most common cases.
The assumption of widelyspaced real poles which leads to (4.126) & (4.127) is tantamount to assuming the Miller capacitance, Cgd1*gm1*R2 >> C2. This is reasonable in the common case where Cgd1 includes a compensation capacitor Cc so that the Miller effect makes it much larger than C2.
Then, in order to arrive at (4.133) from (4.127), we assume C2 >> Cgs1, also reasonable assuming the opamp is loaded by a capacitance much bigger than Cgs1. This then allows us to neglect the term Cgs1*Cgd1 in the denominator of (4.128) as being much smaller than C2*Cgd1.
Looking at equation 1.90 and example 1.13，I find Cov having different definitions.
Yes – you are correct. Example 1.13 should more properly refer to C_{ov}‘, the overlap capacitance normalized per unit length of gate width. Hence, C_{ov} = C_{ov}‘ W. Thanks for catching this!
In EXAMPLE 1.16,I have a question about the definition of Ioff. Only when VGS=0,we have Ioff. Then Ioff has nothing to do with the changes of VGS,and it is a function of vth. Is that Right,Dr Carusone?
Yes – your interpretation is consistent with equation (1.121). This definition is particularly suitable for CMOS digital logic circuits where the MOSFETs act as switches that are OFF when V_{GS} = 0. In analog circuits, sometimes the source (and even the body terminal) may not be at ground. It may be possible to reduce subthreshold leakage in these cases by decreasing V_{GS} even further, below zero. Thanks for raising this confusing point!
The second pole and zero frequency is incorrect in example 4.7 on page 159163. The circuit shows 1pF, but the calculations must have used 100pF, because the frequency is off by 100.
Thanks very much for catching this. It’s now reflected in the errata.
On page 182,Equation4.168 Rout is rds2//RL.I think Rout is gm2rds2rds1//RL;
Equation4.170,gs2=1/rs2=gin2+gds1,gin2 I think is gm2;
Equation4.171 I think gin2 is gm2 too.
Equation (4.168) refers to r_{d2} which is the smallsignal resistance seen looking into the drain of Q_{2}, not simply r_{ds2} as you suggest. The substitution for r_{d2} is then made in (4.174). In equations (4.170) and (4.171), note that g_{in2} is approximately g_{m2}, but is more accurately given by the expression in (4.160) which takes into account the circuit’s finite load impedance.
Page 124, last paragraph, first sentence:
“…, the input impedance, rout, is found to be 1/gm…” > rout should be rin.
Equation 6.71, page 261. It says Veff7 / Veff16 = …
I assume it should be Veff7 / Veff6? (since there’s not even a Q16).
No, not quite, although of course there is a typo there. In fact, in equation (6.71) Veff16 should be replaced by Veff9, the triode device realizing Rc. Thanks.
On page 420, example 10.3.
There’s a print error on Vdd=1V which should be 2V according to the solution below.
Equation 1.146: I think there should be Dp instead of Dn
Page 197 Figure 4.37: Dependent current source arrow direction should be up in order to model + gain of the active load diff. pair.
Page 196 Fig. 4.35 righthand side dependent current source arrow should be upwards since vgs2=vg1/2. Then page 195 Eqn. 4.226 becomes positive.
Figure 7.13 – Based on Ye’s Paper, the terminals of OPAMP are incorrectly connected. Should they not be reversed.
Yes – for Fig. 7.13(b) only they should be reversed in order to provide negative feedback, and in accordance with Ye’s paper. Fig. 7.13(a) is correct as is. Thanks!
Page 510, Ex. 12.10, Fig. 12.41(b): Output differential voltage polarization should be reversed in order to have a negative gain of (1/(sRC)) as stated in the solution.
Page 225 example 4.9 solution:
Fig. 5.13b: simplified smallsignal model of the diff pair’s dependent current source direction should be reversed (Av=+gm*Rout).
Then Fig. 5.13c becomes correct for dependent current source:
gm1(VinVout)”up arrow”
= gm1(0Vt) “up arrow”
=gm1*Vt “up arrow”
= gm1*Vt “down arrow”
Page 170 example 4.8 solution:
In the first sentence, Miller feedback admittance is stated as “Y(s)=1/sC”,which should be Y(s)=sC.
polarity of opamp on fig 7.15 is wrong!!!
on page 446 says Since this charge is negative, it will cause the junction voltages to have negative glitches.
i think positive charge is right since switch of Fig.11.3 is PMOS transistor.
I think you are confused by the symbol in Fig. 11.3. In that figure, Q1 is in fact an NMOS transistor, as indicated by the arrowhead.
In p.145 sec. 4.1 last sentence in last paragraph.
“Further, the real parts of all the poles will be positive for stable transfer functions.”
The term “positive” should be modified as “negative”!
I noticed that on page 148 that with f=10Hz, ω = 2πf = 62.83 rad/s and not 628.3 rad/s
The caption on Fig 4.1 on page 151 says lowpass circuit but the circuit is a highpass.
In fact, the transfer function for Vin/Vout of that circuit is, indeed, lowpass. You may find the analysis in Example 4.3.
The H(s) in the Ex. 4.3 is a highpass filter, sRC/(1+sRC). Which part of the example shows a lowpass transfer function? I got a little confused by this. The basic formula looks like lowpass, but as the circuit transfer function derives, it is clear a highpass filter, doesn’t it? Thank you.
The capacitor acts like open circuit at DC and short circuit to ground at very high frequencies. So the circuit will pass the DC content and the capacitor will attenuate the signal at very high frequencies. This result can also be confirmed with mathematical transfer function of the circuit:
The transfer function is just a voltage divider between the capacitor and the resistor impedances:
H(s) = (1/Cs) / (R+1/Cs) = 1/(1+RCs)
which is a lowpass transfer function. (If you replace s=0 for DC: H=1; and if you replace s=infinity: H=0)
Equation (2.19)
It should be identical with Eq. (2.18)
σ^2(∆VGS) = … = 1/WL[Avt0^2+(ID/gm)^2*(Ak’^2)]